A simple switching power supply on IR2153 (D) for an amplifier and more. Four switching power supplies on IR2153

Hello everybody!

Background:

The site has a circuit for audio frequency power amplifiers (ULF) 125, 250, 500, 1000 watts, I chose the 500 watt option, because in addition to radio electronics, I’m also a little fond of music and therefore I wanted something better from ULF. The circuit on the TDA 7293 did not suit me, so I decided to use the 500 watt field effect transistors. From the beginning, I almost assembled one ULF channel, but work stopped for various reasons (time, money, and the unavailability of some components). As a result, I bought the missing components and finished one channel. Also, after a certain time, I collected the second channel, set it all up and tested it on the power supply from another amplifier, everything worked at the highest level and I liked the quality very much, I didn’t even expect it to be so. Separate, many thanks to the radio amateurs Boris, AndReas, nissan who have collected it all the time, helped in setting it up and in other nuances. Next up was the power supply. Of course, I would like to make a power supply on a conventional transformer, but again, everything stops at the availability of materials for the transformer and their cost. Therefore, I decided to stop at the UPS after all.

Well, now about the UPS itself:

I used IRFP 460 transistors, because I did not find them indicated on the diagram. I had to put the transistors on the contrary by turning 180 degrees, drill more holes for the legs and solder the wires (see the photo). When I made a printed circuit board, I later only realized that I couldn’t find the transistors I needed as in the diagram, I installed those that were (IRFP 460). Transistors and output rectifier diodes must be installed on the heat sink through insulating heat-conducting gaskets, and radiators must also be cooled with a cooler, otherwise transistors and rectifier diodes may overheat, but the heating of transistors of course also depends on the type of transistors used. The lower the internal resistance of the field worker, the less they will heat up.

Also, I have not yet installed a Varistor of 275 Volts at the input, since it is not in the city and I have it too, but it is expensive to order one part via the Internet. I will have separate electrolytes for the output, because they are not available for the required voltage and the size is not suitable. I decided to put 4 electrolytes of 10,000 microfarads * 50 volts, 2 in series per arm, in total, each arm will have 5000 microfarads * 100 volts, which will be completely enough for the power supply, but it is better to put 10,000 microfarads * 100 volts per arm.

The diagram shows the resistor R5 47 kOhm 2 W for powering the microcircuit, it should be replaced with 30 kOhm 5 W (preferably 10 W) in order for the IR2153 chip to have enough current at a heavy load, otherwise it may go into protection against a lack of current or it will pulsate voltage will affect the quality. In the author's circuit, it costs 47 kOhm, which is a lot for such a power supply unit. By the way, the resistor R5 will get very hot, don't worry, the type of these circuits on IR2151, IR2153, IR2155 for power supply is accompanied by a strong heating of R5.

In my case, I used an ETD 49 ferrite core and it was very hard for me to fit onto the board. At a frequency of 56 kHz, according to calculations, it can give up to 1400 watts at this frequency, which in my case has a margin. You can also use a toroidal or other shape of the core, the main thing is that it would be suitable in terms of overall power, permeability and, of course, that there would be enough space to place it on the board.

Winding data for ETD 49: 1 = 20 turns with 0.63 wire in 5 wires (220 volt winding). 2-ka \u003d main power bipolar 2 * 11 turns with a wire 0.63 in 4 wires (winding 2 * 75-80) volts. 3-ka \u003d 2.5 turns with a wire 0.63 in 1 wire (12 volt winding, for soft start). 4-ka \u003d 2 turns with a wire 0.63 in 1 wire (an additional winding for powering preliminary circuits (tone block, etc.). The transformer frame needs a vertical design, I have a horizontal one, so I had to fence it. It can be wound in a frameless design. On other types you will have to calculate the core yourself, you can use the program that I will leave at the end of the article.In my case, I used a bipolar voltage of 2 * 75-80 volts for a 500-watt amplifier, why less, because the load of the amplifier will not be 8 ohms but 4 ohms.

Setup and first run:

When starting the UPS for the first time, be sure to install a 60-100 watt light bulb in the gap between the network cable and the UPS. When you turn it on, if the light does not light up, then it's already good. At the first start, short circuit protection may turn on and the HL1 LED will light up, since high-capacity electrolytes take a huge current at the moment of switching on, if this happens, then you need to twist the multi-turn resistor clockwise until it stops, and then wait until the LED goes out in turned off and try to turn it on again to make sure the UPS is working, and then adjust the protection. If everything is soldered correctly and the correct part ratings are used, the UPS will start. Further, when you make sure that the UPS turns on and there are all voltages at the output, you need to set the protection threshold. When setting up protection, be sure to load the UPS between the two arms of the main output winding (which is for powering the ULF) with a 100-watt light bulb. When the HL1 LED lights up when the UPS is turned on under load (a 100-watt lamp), you need to turn the variable multi-turn resistor R9 2.2 kOhm counterclockwise until the protection is activated when turned on. When the LED lights up when turned on, you need to turn it off and wait until it goes out and gradually twist it clockwise in the off state and turn it on again until the protection stops working,
you just need to turn a little, for example, 1 turn and not immediately by 5-10 turns, i.e. turned it off, turned it on and turned it on, the protection worked - again the same procedure several times until you reach the desired result. When you set the desired threshold, then, in principle, the power supply is ready for use and you can remove the mains voltage light and try to load the power supply with an active load, for example, 500 watts. Of course, you can play around with the protection as you like, but I don’t recommend testing with A short circuit, since this can lead to a malfunction, although there is protection, some capacitance will not have time to discharge, the relay will not respond instantly or it will stick and there may be a nuisance. Although I accidentally and not accidentally made a number of closures, the protection works. But nothing is eternal.

So the first power supply, let's conditionally call it "high-voltage":

The circuit is classic for my switching power supplies. The driver is powered directly from the mains through a resistor, which reduces the power dissipated on this resistor, compared to powering from the +310V bus. This power supply has a soft start (inrush current limit) circuit on the relay. Soft start is powered by a quenching capacitor C2 from a 230V network. This power supply is equipped with protection against short circuit and overload in the secondary circuits. The current sensor in it is the resistor R11, and the current at which the protection is triggered is regulated by the tuning resistor R10. When the protection is triggered, the HL1 LED lights up. This power supply can provide output bipolar voltage up to +/-70V (with these diodes in the secondary circuit of the power supply). The pulse transformer of the power supply has one primary winding of 50 turns and four identical secondary windings of 23 turns. The cross section of the wire and the core of the transformer are selected based on the required power that must be obtained from a particular power supply.

The second power supply, we will conditionally call it a “self-powered UPS”:

This unit has a circuit similar to the previous power supply, but the fundamental difference from the previous power supply is that in this circuit, the driver feeds itself from a separate transformer winding through a quenching resistor. The remaining nodes of the scheme are identical to the previous presented scheme. The output power and output voltage of this unit is limited not only by the parameters of the transformer, and the capabilities of the IR2153 driver, but also by the capabilities of the diodes used in the secondary circuit of the power supply. In my case, this is KD213A. With these diodes, the output voltage cannot be more than 90V, and the output current cannot be more than 2-3A. The output current can be higher only if radiators are used to cool the KD213A diodes. It is worthwhile to additionally dwell on the T2 throttle. This inductor is wound on a common ring core (other types of cores can also be used), with a wire of the section corresponding to the output current. The transformer, as in the previous case, is calculated for the corresponding power using specialized computer programs.

Power supply number three, let's call it "powerful on 460x transistors" or simply "powerful 460":

This scheme is already more significantly different from the previous schemes presented above. There are two main big differences: protection against short circuit and overload is performed here on a current transformer, the second difference is the presence of additional two transistors in front of the keys, which allow isolating the high input capacitance of powerful keys (IRFP460) from the driver output. Another small and not significant difference is that the limiting resistor of the soft start circuit is located not in the +310V bus, as it was in the previous circuits, but in the 230V primary circuit. The circuit also has a snubber connected in parallel with the primary winding of the pulse transformer to improve the quality of the power supply. As in the previous schemes, the protection sensitivity is regulated by a trimmer resistor (in this case, R12), and the HL1 LED signals the protection operation. The current transformer is wound on any small core that you have at hand, the secondary windings are wound with a wire of small diameter 0.2-0.3 mm, two windings of 50 turns each, and the primary winding is one turn of wire sufficient for your output power section.

And the last impulse for today is a “switching power supply for light bulbs”, we will conditionally call it that.

Yes, don't be surprised. Once there was a need to assemble a guitar preamplifier, but the necessary transformer was not at hand, and then this impulse converter, which was built just for that occasion, helped me a lot. The scheme differs from the previous three in its maximum simplicity. The circuit does not have, as such, protection against a short circuit in the load, but there is no need for such protection in this case, since the output current through the secondary bus + 260V is limited by resistor R6, and the output current through the secondary bus + 5V is limited by the internal overload protection circuit of the stabilizer 7805. R1 limits the maximum inrush current and helps cut off mains noise.

When assembling some next device, the question of how to power it is more and more tormenting. Yes, it’s good when there are a lot of different equipment where there are suitable transformers, and if you rewind ??? Rewinding a transformer is not a pleasant task, even if applications for calculating a transformer help in calculations, the rewinding process itself is often annoying.

I remember how it was TSSh-180, a good anode-filament trance, and I had to rewind. I probably wound it for two days, plus shed it with varnish so that the insulation would be better and not buzz ... I collected it, such a healthy one. Himself weighing 3 kg and almost fell on his leg. I thought about it all and decided to switch to switching power supplies and there are a lot of reasons for this.

Reasons for choosing switching power supplies:

1. P The first and not unimportant reason is financial. Here we have the same TSSh-180 a.-incandescent costs 150-180 UAH. While the assembled SMPS 200W on IR2153 will cost 130-160 UAH. Yes, the difference is not great, but your house is full of the necessary details. For example, I bought only IRF740 and IR2153 and paid 40 UAH. How's the difference?? And I also got rid of the trash a little)) And it’s also unforgettable that the bridge and the banks are already in the calculation, and you also need to buy this for the trance. And good jars about how well they cost. And on the SMPS instead of 22,000mF, you can put 3300mF and you won’t even notice the difference in filtering

2. In The second reason is the size. The trances are heavy, so 200 watts weighing 3-4 kg, it is replaced by an SMPS with a mass of 300 g and a board size of about 120 * 120 mm. It is convenient to assemble something powerful in a DVD box, for example Lanzar ...

3. E then a low level of interference in the range of 20-20,000 Hz. This is very good for a low-frequency amplifier, even excellent. No interference, no background.

On the diagram we see the power part in which there is: protective circuits (R1, R2, FU1) filter C-R-C (C1, L1, C1), rectifier with filter divider (VD1 (400V 3A), C3, C4, C6, C7, R44, R6) and a key part which includes two mosfets (VT1, VT2), a transformer (T1) and two noise suppression circuits (R8C9, C8R7)

Nothing complicated in the control part. The supply part of the microcircuit consists of a ballast resistor R9, a zener diode VD2. filter C10C11, and another ballast resistor R10. In the course of work, you may have to pick up R9R10.
The PWM frequency is set by R11C13. And it is calculated according to the formula f = 1 / 1.4 * (R11 + 75 Ohm) * C13. In our case f=1/1.4*(10000+75)*0.000000001=70896 Hz= 70.9 kHz. Be careful with toes

Well, there really is nothing to tell: Dual diode VD4, filter-rectifier C14-L3-C15-C16 and that's it. Remember when calculating that this is not a stabilized PSU and the voltage can float. Therefore, it is better to enter a couple of volts less when calculating

The application for calculating the pulse transformers will help you with the calculation of the transformer. Advice to wind the secondary with a scythe of a thinner wire in order to avoid the skin effect.

By the way, one of my friends from such a circuit is powered by 2.1 assembled on a TDA2030A with a total power of 65W. This is a small part of what the SMPS produces on the IR2153, but it has been working for a year. Yes, again, a 70W transformer right now costs the same as the SMPS unit on IR2153, so the SMPS also has a reserve of 130W ...

That's all, thank you all for your attention and good luck with the assembly ...

About the article.
There are many schemes in the global garbage dump using this microcircuit and the description, do this and that ... But how and why? Will it work? The answer to the last question is often no! There are a lot of "Miracle" seals and tips to use a 1000uF x500V capacitor, which cannot be found or will cost half the salary.
I will try to describe what I had to face when building the device, how it was decided, to reduce everything to simple and understandable principles, using which everyone can decide what he needs.

About the "irka" itself - IR2153.
The microcircuit is designed for use in electronic ballasts of economical lamps, these are microscopic power devices, operate at frequencies of the order of 30 kHz, and do not have specially provided protection and control circuits. This gives food for thought!
The IR2153 is low power and can be powered simply through a pull-down resistor, and has a split for the upper and lower half-bridge switches, so there is no need to wind transformers or use optical separation of the switch control signals.
This makes the chip attractive not only for amateurs, but also for serious brands that produce products in series!

And so, the project itself.

The goal was to build a simple, as versatile as possible, power supply module with a power of about 200W.
The scope of application is from powering halogen lamps to UMZCH, etc. , oddly enough, in terms of the cost of materials, this module can compete with factory transformers for halogen lamps, even more so in other areas of application.

Accessories:
C1 - 220uF x 450V (everything is modest with us 🙂)
C2, C10 - 1uF x 400V, film
C3 - 470..1000uF x 25V
C4,C5,C9,C8 - 0.22..0.47uF x 63V ceramic (or film)
C6 - 10uF x 63V
C7 -1nF ceramic setting the oscillator frequency.
R1,R2 - 65K 2W
R3 - 12K, setting the frequency of the generator.
R4 - 8.2K
VD1-UF4007
VT1,VT3 - BC640
VT2,VT4 - BC639
T1,T2 - IRF840
DIL8-IR2153
F2-Fast 2A

Setting.

Before the first start, the circuit is not assembled completely, the "upper" key T1 is not soldered.
When the device is connected to the network, voltages appear:
on capacitors C1 and C2 about 300V, on C3 and C4 14..15V (1 and 4 pins of IR2153), on C5 and C6 - 14..15V (8 and 6 pins of IR2153).
at the outputs of the IR2153 driver, between pins 4 and 5 - 12..14V, between pins 6 and 7 - 12..14V, these voltages must be equal.
at the outputs of the driver amplifier, between the COM and OUT points (lower key driver) - 11..13V, between the VS and OUT points (upper key driver) - 11..13V, these voltages must be equal.
All these voltages are measured with a multimeter in DC voltage measurement mode, at the limit of 750 \ 1000V, for greater safety and so that tenths and hundredths of a volt do not fool your head.
If possible, then you can check the signals at the outputs of the IR2153 and the driver amplifier with an oscilloscope.
Attention!!
If the measurement points are mixed up or shorted, something or everything will burn!
The oscilloscope ground must NOT be grounded.
After a successful check, you can solder the top key T1.

Nodes of the converter, the principle of their work.

Network rectifier.

Capacitor C1 is chosen with a relatively small capacity, as it is enough for the operation of the unit, if the unit is used to power halogen lamps, then it is enough, if you power the UMZCH and other devices, then additional filtering from the mains background of 100 Hz is easily carried out after transformation and rectification, with low-voltage electrolytes, this is both better and cheaper, since the cost of a 10000mkFx35V capacitor is much lower than the cost of a 220mkFx450V capacitor.

The small capacitance of the mains rectifier does not affect the operation of the IR2153, since it has its own zener diode (built-in) and a filter and is powered normally, and the switches, in the worst case, will only transmit 100 Hz ripples through the transformer.

Capacitor C2 plays an important role in the rectifier, it handles rapidly changing voltages that a slow electrolytic capacitor cannot handle.
Capacitor C2 blocks RF interference on the power buses, thanks to it the circuit can output normal pulses, thanks to it the recuperation system can work normally, which reduces voltage surges on transistors, improves the reliability and quality of the circuit.
Very often they "forget" to put it.

IR2153 - Power.

When IR2153 is powered through a quenching resistor (R1, R2), there is a danger of a voltage drop to critical values, when this resistance decreases, the driver’s power improves, but the heating of the board and the entire device increases.

Things like: an additional external driver, increasing the conversion frequency, increasing the gate capacitance (increasing the power of the output transistors), and even simply increasing the current consumption of the load, increase the consumption from the IR2153 power supply. The only salvation is a significant current margin for the IR2153 power supply.
Alternative power supply methods: from an additional 15V power supply, with capacitor decoupling from the 6th output of the microcircuit (this is also a half-bridge output), from an additional transformer winding.

For this scheme, for example, the power situation looks like this:
Conversion frequency 50KHz, IRF840, quenching resistor in the power supply 2x 65K 2W (32K 4W).
In the circuit, only the lower fet - 15.9V
Both fetas are connected - 12.3V
Connected transformer - 13V

I lowered the conversion frequency to just something up to 40KHz !!
Under load 100W - voltage 14V

Obviously, up to 100W of consumption at a conversion frequency of 60KHz, it was possible to get by with power from a 32K \ 4W resistor, with more power there is no way.
To ensure high-quality power supply to the driver, I made an additional 25V winding on the transformer and applied voltage through 2 100Ω resistors to the rectifier bridge, the diode bridge (UF4007 x4) is soldered in place of the input power terminals. How this is done can be seen in Photo 1.

IR2153 - Generator frequency.

Pauses between pulses (Dead time) for this controller are fixed at 1.2 µs, because of this, with increasing frequency, the duty cycle of the pulses decreases.
So, for a frequency of 50 kHz, the pauses are 12%, for 100 kHz, all 24%.
With increasing frequency, the bandwidth of most ferrites improves, but the filling of the pulses decreases.

Brief information about one of the characters. Miller effect.
Its reason is the capacitance between the input of the electrical cascade and its output, it does not depend on the layout of the printed circuit board and even on the cascade circuit, for example, it was discovered by Miller in a tube amplifier and still haunts electronics everywhere.
In converter circuits, the Miller effect creates through currents and heating of transistors at idle, when the converter is operating on a heavy load, it threatens to disable the unit.

The IR2153 has its own built in driver and the first time I ran the unit using it.
Here's how it works.
IRF840 gate signal:

IRFP460 gate signal:

Oscillograms are distinguished by a very smooth front (rising of the pulse), isn't it?
Even if the pulse frequency is reduced to 30 kHz, there are huge drafts from the Miller effect, these current poke can be admired on the oscillograms.
Nevertheless, you can hear, or rather read, that this is how everything works !! This can be trusted, the circuit will probably not immediately light up, especially on a large radiator.

The driver is very weak (200mA per pulse), designed for low power transistors, because this is a microcircuit for ballast in lamps!
The transistor repeater driver used in this project greatly improves the situation.

Signal with transistor follower:

The external driver reduces the Miller effect, increases the efficiency of the unit.
All these waveforms were with a completely empty half-bridge output, no snubbers, not even a transformer winding.
Now signals from loaded transistors.
IRF840 load 200W (signal is smooth and transistors don't heat up much):

IRF840 10KV transformer + lds

Another moment, when at least the primary winding of the transformer is connected, the transistors stop heating up and Miller's pokes disappear along the front, they disappear, but Miller does not disappear anywhere, and here it appears again, now by the decay of the pulse, on the oscillograms from the unit under load! Voila! And it is clear that even a powerful external driver could hardly keep the unit from a fire. So a driver is needed to improve the reliability of the unit.
The cost of the given driver is only 10% of the cost of IR2153.

While he was shredding the block, he assembled another driver, it crushes Miller even better, although the transistors are still the same, apparently due to the increase in the gain of the cascade, during tests he simply cut the existing driver seal and soldered the transistor. Scheme and waveform, block at idle:

Transformer(s).

At its core, a pulse transformer for forward circuits is no different from a conventional 50Hz AC transformer.
At idle, the current through the primary winding, determined by the inductive reactance, is very small, and should be just that.
A loaded transformer transforms the resistance of the load connected to the secondary winding, in accordance with the transformation ratio (ratio of turns of the primary and secondary windings) and the current in the primary winding is determined by the transformed load resistance.

The thickness of the wires is determined by the maximum current, and the design of the winding, with multilayer, the wire is needed thicker.
A core with an increase in frequency transmits energy better, but remagnetization losses can increase in it, with a decrease in frequency, the ferrite enters saturation more easily, which can cause a sharp decrease in the inductance of the primary winding by thousands of times and the unit burns out.

An example of a "folk" transformer, for a half-bridge 50..60KHz.
Ferrite grade 2000NMS, from a line transformer TVS110pts15, primary winding 150V - 30..40 turns of wire, secondary is calculated for the desired voltage, based on the required voltage and the volt / turn coefficient in the primary winding.
For example for this core:
The power supply of the output stage is a half-bridge 310V, then the voltage of the pulses on the primary winding of the transformer is 150V
Primary winding at 150V - 30 turns (5V / turn)
Secondary winding at 15V - 3 turns

If the secondary winding has a small number of turns and poor filling of the transformer window, then you can wind the secondary winding with several parallel conductors, which are then soldered in parallel, so you can reduce the heating of the secondary winding and increase the magnetic coupling of the windings. For one such core, the bandwidth is approximately 500W, and if necessary, the cores can be paralleled, proportionally reducing the number of turns of the primary winding, so for two cores you can take 20 turns, for three - 15 turns.

The design of such a transformer is certainly not optimal, but it is easy to make it at home and by winding the primary and secondary windings on different sides of the ferrite, you can achieve a soft connection between the windings, which can save the device in case of a short circuit in the secondary winding.

Transformer from this project.
Core made of 8 rings TN2010-3E25, 5340nH (20.6x9.2x7.5mm)
Primary winding 150V - 12 Turns of wire in PVC insulation
Secondary winding - 1 turn
Here, the weak link is the core material, suitable only for weak magnetic fields, can easily saturate and burn the power supply. But in principle, the design is promising for amateurs, only choose a different material.

I hope that the proposed material will help interested parties to decide on the necessary circuitry to create a device for their needs.
Eee.. remember any unexpected sneeze or unsoldered gate of the transistor will cause an instant and merciless PUSH, because all nodes are galvanically connected, nothing will be saved.

Hello everybody!

Background:

The site has a circuit for audio frequency power amplifiers (ULF) 125, 250, 500, 1000 watts, I chose the 500 watt option, because in addition to radio electronics, I’m also a little fond of music and therefore I wanted something better from ULF. The circuit on the TDA 7293 did not suit me, so I decided to use the 500 watt field effect transistors. From the beginning, I almost assembled one ULF channel, but work stopped for various reasons (time, money, and the unavailability of some components). As a result, I bought the missing components and finished one channel. Also, after a certain time, I collected the second channel, set it all up and tested it on the power supply from another amplifier, everything worked at the highest level and I liked the quality very much, I didn’t even expect it to be so. Separate, many thanks to the radio amateurs Boris, AndReas, nissan, who have collected it all the time so far, helped in setting it up and in other nuances. Next up was the power supply. Of course, I would like to make a power supply on a conventional transformer, but again, everything stops at the availability of materials for the transformer and their cost. Therefore, I decided to stop at the UPS after all.

Well, now about the UPS itself:

Setup and first run:

Measurements after assembling the UPS:

Measurements between shoulders:
U in - 225 volts, load - 100 watts, U out + - = 164 volts
U in - 225 volts, load - 500 watts, U out + - = 149 volts
U in - 225 volts, load - 834 watts, U out + - = 146 volts

There is a sitting, of course. With a load of 834 watts in front of the input rectifier, the voltage sags from 225 volts to 220 volts, after the rectifier it sags by as much as 20 volts from 304 volts to 284 volts at a load of 834 watts. But in principle, the output sag on each arm is 9 volts, which in principle is acceptable, since the UPS is not stabilized.

Thank you all for your attention.

PULSE POWER SUPPLY WITH YOUR HANDS ON IR2153

Functionally, the IR2153 microcircuits differ only in the diode installed in the planar package.

Functional diagram of IR2153

Functional diagram of IR2153D

To begin with, let's look at how the microcircuit itself works, and only then we will decide which power supply to assemble from it. First, let's look at how the generator itself works. The figure below shows a fragment of a resistive divider, three op-amps and an RS flip-flop:

At the initial moment of time, when the supply voltage was just applied, the capacitor C1 is not charged at all the inverting inputs of the op-amp, there is zero, and at the non-inverting positive voltage generated by the resistive divider. As a result, it turns out that the voltage at the inverting inputs is less than at the non-inverting ones, and all three op-amps at their outputs form a voltage close to the supply voltage, i.e. log unit.
Since the input R (setting zero) on the trigger is inverting, then for it it will be a state in which it does not affect the state of the trigger, but at the input S there will be a one log, which also sets a log one at the trigger output and the capacitor Ct through the resistor R1 will start charging. On the image voltage across Ct shown as blue line,red - voltage at the output DA1, green - at the output DA2, A pink - at the RS trigger output:

As soon as the voltage at Ct exceeds 5 V, a log zero is formed at the output of DA2, and when, continuing to charge Ct, the voltage reaches a value slightly more than 10 volts, a log zero will appear at the output of DA1, which in turn will set the RS trigger to a log zero state. From this moment, Ct will start to discharge, also through the resistor R1, and as soon as the voltage across it becomes slightly less than the set value of 10 V, a log unit will appear again at the DA1 output. When the voltage on the capacitor Ct becomes less than 5 V, a log unit will appear at the output of DA2 and turn the RS flip-flop to the state of one and Ct will start charging again. Of course, at the inverted output RS of the flip-flop, the voltage will have opposite logical values.
Thus, at the outputs of the RS trigger, opposite in phase, but equal in duration, log one and zero levels are formed:

Since the duration of the control pulses IR2153 depends on the charge-discharge rate of the capacitor Ct, it is necessary to carefully pay attention to flushing the board from the flux - there should not be any leaks from either the capacitor terminals or the printed circuit conductors of the board, since this is fraught with magnetization of the power transformer core and failure power transistors.
There are also two more modules in the microcircuit - UV DETECT And LOGIK. The first of them is responsible for the start-stop of the generator process, depending on the supply voltage, and the second generates pulses DEAD TIME, which are necessary to exclude the through current of the power stage.
Then there is a separation of logical levels - one becomes the control upper arm of the half-bridge, and the second the lower one. The difference lies in the fact that the upper arm is controlled by two field-effect transistors, which, in turn, control the final stage "torn off" from the ground and "torn off" from the supply voltage. If we consider a simplified circuit diagram of the inclusion of IR2153, then it turns out something like this:

Pins 8, 7 and 6 of the IR2153 chip are the outputs VB, HO and VS, respectively, i.e. high-side control power supply, the output of the high-side control final stage, and the negative wire of the high-side control module. Attention should be paid to the fact that at the moment of switching on, the control voltage is present at the Q RS of the flip-flop, therefore the low-side power transistor is open. Capacitor C3 is charged through diode VD1, since its lower output is connected to a common wire through transistor VT2.
As soon as the RS trigger of the microcircuit changes its state, VT2 closes, and the control voltage at pin 7 of the IR2153 opens the transistor VT1. At this point, the voltage at pin 6 of the microcircuit begins to increase, and to keep VT1 open, the voltage at its gate must be greater than at the source. Since the resistance of an open transistor is equal to tenths of an ohm, the voltage at its drain is not much greater than at the source. It turns out that keeping the transistor in the open state requires a voltage of at least 5 volts more than the supply voltage, and it really is - the capacitor C3 is charged up to 15 volts and it is he who allows you to keep VT1 in the open state, since the energy stored in it in this the moment of time is the supply voltage for the upper arm of the window stage of the microcircuit. Diode VD1 at this point in time does not allow C3 to be discharged to the power bus of the microcircuit itself.
As soon as the control pulse at pin 7 ends, the transistor VT1 closes and then VT2 opens, which again recharges the capacitor C3 to a voltage of 15 V.

Quite often, amateurs install an electrolytic capacitor with a capacity of 10 to 100 microfarads in parallel with capacitor C3, without even delving into the need for this capacitor. The fact is that the microcircuit is capable of operating at frequencies from 10 Hz to 300 kHz and the need for this electrolyte is relevant only up to frequencies of 10 kHz, and then, provided that the electrolytic capacitor is of the WL or WZ series, they technologically have a small ers and are better known as computer capacitors with inscriptions in gold or silver paint:

For popular conversion frequencies used in the creation of switching power supplies, frequencies are taken above 40 kHz, and sometimes adjusted to 60-80 kHz, so the relevance of using an electrolyte simply disappears - even a capacitance of 0.22 uF is already enough to open and hold the SPW47N60C3 transistor in the open state , which has a gate capacitance of 6800 pF. To calm my conscience, a 1 uF capacitor is placed, and giving an amendment to the fact that IR2153 cannot switch such powerful transistors directly, then the accumulated energy of capacitor C3 is enough to control transistors with a gate capacity of up to 2000 pF, i.e. all transistors with a maximum current of about 10 A (the list of transistors is below in the table). If you still have doubts, then instead of the recommended 1 uF, use a 4.7 uF ceramic capacitor, but this is pointless:

It would not be fair not to note that the IR2153 chip has analogues, i.e. microchips with similar functionality. These are IR2151 and IR2155. For clarity, we will summarize the main parameters in a table, and only then we will figure out which of them is better to cook:

CHIP	Maximum driver voltage	Start supply voltage	Stop supply voltage	Maximum current for driving the gates of power transistors / rise time	Maximum current for discharging the gates of power transistors / fall time	Internal zener voltage
				100 mA / 80...120 nS	210 mA / 40...70 nS
				NOT SPECIFIED / 80...150 nS	NOT SPECIFIED / 45...100 nS
				210 mA / 80...120 nS	420 mA / 40...70 nS

As can be seen from the table, the differences between the microcircuits are not very large - all three have the same shunt zener diode for power supply, the start and stop supply voltages for all three are almost the same. The difference lies only in the maximum current of the final stage, which determines which power transistors and at what frequencies the microcircuits can control. Strange as it may seem, but the most hyped IR2153 turned out to be neither fish nor meat - it does not have a normalized maximum current of the last driver stage, and the rise-fall time is somewhat prolonged. They also differ in cost - IR2153 is the cheapest, but IR2155 is the most expensive.
The generator frequency, it is the conversion frequency ( no need to divide by 2) for IR2151 and IR2155 is determined by the formulas below, and the frequency of IR2153 can be determined from the graph:

In order to find out which transistors can be controlled by the IR2151, IR2153 and IR2155 microcircuits, you should know the parameters of these transistors. Of greatest interest when docking a microcircuit and power transistors is the gate energy Qg, since it is it that will affect the instantaneous values of the maximum current of the microcircuit drivers, which means that a table with transistor parameters is required. Here SPECIAL attention should be paid to the manufacturer, since this parameter varies from manufacturer to manufacturer. This is most clearly seen in the example of the IRFP450 transistor.
I understand perfectly well that for a one-time production of a power supply unit, ten to twenty transistors are still a bit too much, nevertheless, I posted a link for each type of transistor - I usually buy there. So click, see prices, compare with retail and the likelihood of buying a leftist. Of course, I'm not saying that Ali has only honest sellers and all goods of the highest quality - there are a lot of crooks everywhere. However, if you order transistors that are manufactured directly in China, it is much more difficult to run into shit. And it is for this reason that I prefer STP and STW transistors, and I don’t even disdain buying from disassembly, i.e. BOO.

POPULAR TRANSISTORS FOR SWITCHED POWER SUPPLY
NAME	VOLTAGE	POWER	CAPACITY SHUTTER	Qg (MANUFACTURER)
NETWORK (220 V)
				17...23nC ( ST)
				38...50nC ( ST)
				35...40nC ( ST)
				39...50nC ( ST)
				46nC ( ST)
				50...70nC ( ST)
				75nC( ST)
				84nC ( ST)


				65nC ( ST)



				46nC ( ST)
				50...70nC ( ST)
				75nC( ST)
				65nC ( ST)
STP20NM60FP				54nC ( ST)






				150nC (IR) 75nC( ST)



				150...200nC (IN)
				252...320nC (IN)
				87...117nC ( ST)

I g \u003d Q g / t on \u003d 63 x 10 -9 / 120 x 10 -9 \u003d 0.525 (A) (1)

With the amplitude of the control voltage pulses at the gate Ug = 15 V, the sum of the output resistance of the driver and the resistance of the limiting resistor should not exceed:

R max = U g / I g = 15 / 0.525 = 29 (ohm) (2)

We calculate the output output impedance of the driver stage for the IR2155 chip:

R on \u003d U cc / I max \u003d 15V / 210mA \u003d 71.43 ohms
R off \u003d U cc / I max \u003d 15V / 420mA \u003d 33.71 ohms

Taking into account the calculated value according to the formula (2) Rmax = 29 Ohm, we come to the conclusion that with the IR2155 driver it is impossible to obtain the specified speed of the IRF840 transistor. If a resistor Rg = 22 Ohm is installed in the gate circuit, we determine the turn-on time of the transistor as follows:

RE on = R on + R gate, where RE - total resistance, R R gate - resistance installed in the gate circuit of the power transistor = 71.43 + 22 = 93.43 ohms;
I on \u003d U g / RE on, where I on is the opening current, U g - gate control voltage value = 15 / 93.43 = 160mA;
t on \u003d Q g / I on \u003d 63 x 10-9 / 0.16 \u003d 392nS
The turn-off time can be calculated using the same formulas:
RE off = R out + R gate, where RE - total resistance, R out - driver output impedance, R gate - resistance installed in the gate circuit of the power transistor = 36.71 + 22 = 57.71 ohms;
I off \u003d U g / RE off, where I off - opening current, U g - gate control voltage value = 15 / 58 = 259mA;
t off \u003d Q g / I off \u003d 63 x 10-9 / 0.26 \u003d 242nS
To the resulting values, it is necessary to add the time of its own opening - closing of the transistor, as a result of which the real time t on will be 392 + 40 = 432nS, and t off 242 + 80 = 322nS.
Now it remains to make sure that one power transistor has time to completely close before the second one starts to open. To do this, add t on and t off getting 432 + 322 = 754 nS, i.e. 0.754µS. What is it for? The fact is that any of the microcircuits, be it IR2151, or IR2153, or IR2155, has a fixed value DEAD TIME, which is 1.2 µS and does not depend on the frequency of the master oscillator. The datasheet mentions that Deadtime (typ.) is 1.2 µs, but there is also a very embarrassing figure from which the conclusion suggests itself that DEAD TIME is 10% of the duration of the control pulse:

To dispel doubts, the microcircuit was turned on and a two-channel oscilloscope was connected to it:

The power supply was 15 V, and the frequency was 96 kHz. As can be seen from the photograph, with a sweep of 1 µS, the duration of the pause is quite a bit more than one division, which exactly corresponds to approximately 1.2 µS. Next, reduce the frequency and see the following:

As you can see from the photo at 47kHz, the pause time didn't really change, hence the sign that says Deadtime (typ.) 1.2 µs is true.
Since the microcircuit was already working, it was impossible to resist one more experiment - to reduce the supply voltage to make sure that the generator frequency increased. The result is the following picture:

However, the expectations were not justified - instead of increasing the frequency, it decreased, and by less than 2%, which can generally be neglected and it should be noted that the IR2153 chip keeps the frequency fairly stable - the supply voltage has changed by more than 30%. It should also be noted that the pause time has slightly increased. This fact is somewhat pleasing - with a decrease in the control voltage, the opening time - closing of the power transistors slightly increases and an increase in the pause in this case will be very useful.
It was also found out that UV DETECT copes with its function perfectly - with a further decrease in the supply voltage, the generator stopped, and with an increase, the microcircuit started up again.
Now let's return to our mathematics, according to the results of which we found out that with 22 Ohm resistors installed in the gates, the closing and opening times are 0.754 µS for the IRF840 transistor, which is less than the 1.2 µS pause given by the microcircuit itself.
Thus, with an IR2155 microcircuit through 22 Ohm resistors, it can quite normally control the IRF840, but the IR2151 will most likely die for a long time, since we needed a current of 259 mA and 160 mA, respectively, to close and open the transistors, and its maximum values are 210 mA and 100 ma. Of course, you can increase the resistances installed in the gates of power transistors, but in this case there is a risk of going beyond DEAD TIME. In order not to engage in fortune-telling on coffee grounds, a table was compiled in EXCEL, which you can take. It is assumed that the supply voltage of the microcircuit is 15 V.
To reduce switching noise and to slightly reduce the closing time of power transistors in switching power supplies, either a power transistor is shunted with a resistor and a capacitor connected in series, or the power transformer itself is shunted in the same circuit. This node is called a snubber. The snubber circuit resistor is chosen with a value of 5–10 times the drain resistance - the source of the field-effect transistor in the open state. The capacitance of the circuit capacitor is determined from the expression:
C \u003d tdt / 30 x R
where tdt is the pause time for switching the upper and lower transistors. Based on the fact that the duration of the transient, equal to 3RC, should be 10 times less than the duration of the dead time value tdt.
Damping delays the opening and closing moments of the field-effect transistor relative to the control voltage drops at its gate and reduces the rate of voltage change between the drain and the gate. As a result, the peak values of the current pulses are smaller, and their duration is longer. Almost without changing the turn-on time, the damping circuit significantly reduces the turn-off time of the field-effect transistor and limits the spectrum of radio interference generated.

With the theory sorted out a bit, you can proceed to practical schemes.
The simplest IR2153 switching power supply circuit is an electronic transformer with a minimum of functions:

There are no additional functions in the circuit, and the secondary bipolar power supply is formed by two rectifiers with a midpoint and a pair of dual Schottky diodes. The capacitance of the capacitor C3 is determined on the basis of 1 microfarad of capacitance per 1 W of load. Capacitors C7 and C8 are of equal capacity and are located in the range from 1 uF to 2.2 uF. The power depends on the core used and the maximum current of the power transistors and theoretically can reach 1500 watts. However, this is only THEORETICALLY , based on the fact that 155 V AC is applied to the transformer, and the maximum current of the STP10NK60Z reaches 10A. In practice, in all datasheets, a decrease in the maximum current is indicated depending on the temperature of the transistor crystal, and for the STP10NK60Z transistor, the maximum current is 10 A at a crystal temperature of 25 degrees Celsius. At a crystal temperature of 100 degrees Celsius, the maximum current is already 5.7 A, and we are talking about the temperature of the crystal, and not the heat sink flange, and even more so about the temperature of the radiator.
Therefore, the maximum power should be selected based on the maximum current of the transistor divided by 3 if this is a power supply for a power amplifier and divided by 4 if this is a power supply for a constant load, such as incandescent lamps.
Given the above, we get that for a power amplifier you can get a switching power supply with a power of 10/3 = 3.3A, 3.3A x 155V = 511W. For a constant load, we get a power supply 10/4 = 2.5 A, 2.5 A x 155V = 387W. In both cases, 100% efficiency is used, which does not happen in nature.. In addition, if we proceed from the fact that 1 μF of the primary power capacitance per 1 W of load power, then we need a capacitor or capacitors with a capacity of 1500 μF, and such a capacitance already needs to be charged through soft start systems.
A switching power supply with overload protection and soft start for secondary power is shown in the following diagram:

First of all, this power supply has overload protection, made on the current transformer. Details on the calculation of the current transformer can be read. However, in the vast majority of cases, a ferrite ring with a diameter of 12 ... 16 mm is quite sufficient, on which about 60 ... 80 turns are wound into two wires. Diameter 0.1...0.15 mm. Then the beginning of one winding is connected to the ends of the second. This is the secondary winding. The primary winding contains one or two, sometimes one and a half turns are more convenient.
Also in the circuit, the values of the resistor R4 and R6 are reduced in order to expand the range of the primary supply voltage (180 ... 240V). In order not to overload the zener diode installed in the microcircuit, the circuit has a separate zener diode with a power of 1.3 W at 15 V.
In addition, a soft start for secondary power was introduced into the power supply, which made it possible to increase the capacity of the secondary power filters to 1000 μF at an output voltage of ±80 V. Without this system, the power supply went into protection at the moment of switching on. The principle of operation of the protection is based on the operation of the IR2153 at an increased frequency at the time of switching on. This causes losses in the transformer and it is not able to deliver maximum power to the load. As soon as the generation through the divider R8-R9, the voltage supplied to the transformer enters the detector VD5 and VD7 and the charging of the capacitor C7 begins. As soon as the voltage becomes sufficient to open VT1, C3 is connected to the frequency-setting chain of the microcircuit and the microcircuit reaches the operating frequency.
Additional inductances for the primary and secondary voltages have also been introduced. The primary power inductance reduces the interference generated by the power supply and goes to the 220V network, and the secondary one reduces RF ripple at the load.
In this version, there are two more additional secondary power supplies. The first is designed to power a computer twelve-volt cooler, and the second is to power the preliminary stages of the power amplifier.
Another sub-variant of the circuit is a switching power supply with a unipolar output voltage:

Of course, the secondary winding counts on the voltage that is needed. The power supply can be soldered on the same board without mounting elements that are not on the diagram.

The next version of the switching power supply is capable of delivering about 1500 W to the load and contains soft start systems for both primary and secondary power, has overload protection and voltage for the forced cooling cooler. The problem of controlling powerful power transistors is solved by using emitter followers on transistors VT1 and VT2, which discharge the gate capacitance of powerful transistors through themselves:

Such forcing the closing of power transistors allows the use of quite powerful instances, such as IRFPS37N50A, SPW35N60C3, not to mention IRFP360 and IRFP460.
At the moment of switching on, the voltage to the primary power diode bridge is supplied through the resistor R1, since the contacts of the relay K1 are open. Further, the voltage, through R5, is supplied to the microcircuit and through R11 and R12 to the output of the relay winding. However, the voltage increases gradually - C10 is quite large capacity. From the second winding of the relay, voltage is supplied to the zener diode and thyristor VS2. As soon as the voltage reaches 13 V, it will already be enough to open VS2 after passing the 12 volt zener diode. It should be recalled here that IR2155 starts at a supply voltage of approximately 9 V, therefore, at the time of opening VS2 through IR2155 it will already generate control pulses, only they will enter the primary winding through resistor R17 and capacitor C14, since the second group of contacts of relay K1 is also open . This will significantly limit the charge current of the secondary power filter capacitors. As soon as the VS2 thyristor opens, voltage will be applied to the relay winding and both contact groups will close. The first shunts the current-limiting resistor R1, and the second shunts R17 and C14.
The power transformer has a service winding and a rectifier based on VD10 and VD11 diodes, from which the relay will be powered, as well as additional feeding of the microcircuit. R14 serves to limit the current of the forced cooling fan.
Used thyristors VS1 and VS2 - MCR100-8 or similar in TO-92 package
Well, at the end of this page, another circuit is all on the same IR2155, but this time it will act as a voltage regulator:

As in the previous version, the power transistors are closed by bipolars VT4 and VT5. The circuit is equipped with a secondary voltage soft start on VT1. The start is made from the vehicle's on-board network, and then the power is supplied by a stabilized voltage of 15 V, fed by diodes VD8, VD9, resistor R10 and zener diode VD6.
In this scheme, there is another rather interesting element - tC. This is a heatsink overheating protection that can be used with almost any inverter. It was not possible to find an unambiguous name, in common people this is a self-resetting thermal fuse, in price lists it usually has the designation KSD301. It is used in many household electrical appliances as a protective or temperature regulating element, since they are produced with different response temperatures. The fuse looks like this:

As soon as the heatsink temperature reaches the cut-out limit of the fuse, the control voltage from the REM point will be removed and the inverter will turn off. After the temperature drops by 5-10 degrees, the fuse will be restored and supply control voltage and the converter will start up again. The same thermal fuse, well, or a thermal relay can also be used in network power supplies by controlling the temperature of the radiator and turning off the power, preferably low-voltage, going to the microcircuit - the thermal relay will work longer this way. You can buy KSD301.
VD4, VD5 - fast diodes from the SF16, HER106 series, etc.
Overload protection can be introduced into the circuit, but during its development, the main emphasis was on miniaturization - even the softstart node was a big question.
The manufacture of winding parts and printed circuit boards are described on the following pages of the article.

Well, in the end, several circuits of switching power supplies found on the Internet.
Scheme No. 6 is taken from the SOLDERING IRON website:

In the next power supply on the self-clocked driver IR2153, the capacity of the booster capacitor is reduced to a minimum sufficiency of 0.22 microfarads (C10). The microcircuit is powered from the artificial midpoint of the power transformer, which is not important. There is no overload protection, the shape of the voltage supplied to the power transformer is slightly corrected by the inductance L1:

Choosing schemes for this article, I came across this one. The idea is to use two IR2153s in a bridge converter. The idea of the author is quite understandable - the output RS of the trigger is fed to the input Ct and, logically, control pulses opposite in phase should be formed at the outputs of the slave microcircuit.
The idea intrigued and an investigative experiment on the topic of working capacity testing was carried out. It was not possible to get stable control pulses at the outputs of IC2 - either the upper driver was working, or the lower one. In addition, the pause phase DEAD TIME, on one chip relative to another, which will significantly reduce the efficiency and the idea was forced to be abandoned.

A distinctive feature of the next power supply on the IR2153 is that if it works, then this work is akin to a powder keg. First of all, an additional winding on the power transformer to power the IR2153 itself caught my eye. However, there is no current-limiting resistor after diodes D3 and D6, which means that the fifteen-volt zener diode inside the microcircuit will be VERY heavily loaded. What happens when it overheats and thermal breakdown can only be guessed at.
Overload protection on VT3 shunts the time-setting capacitor C13, which is quite acceptable.

The last acceptable power supply circuit on the IR2153 is nothing unique. True, the author for some reason too much reduced the resistance of the resistors in the gates of power transistors and installed zener diodes D2 and D3, the purpose of which is not very clear. In addition, the capacitance C11 is too small, although it is possible that we are talking about a resonant converter.

There is another option for a switching power supply using IR2155 and it is for controlling a bridge converter. But there, the microcircuit controls power transistors through an additional driver and a matching transformer, and we are talking about induction melting of metals, so this option deserves a separate page, and everyone who understands at least half of what they read should go to the page with printed circuit boards.

VIDEO INSTRUCTIONS FOR SELF-ASSEMBLY
PULSE POWER SUPPLY BASED ON IR2153 OR IR2155

A few words about the manufacture of pulse transformers:

How to determine the number of turns without knowing the brand of ferrite:

Electronic ballasts. Simple electronic ballast on the IR2153 chip

Consider a simple electronic ballast circuit based on the IR2153 (IR2151) microcircuit, shown in fig. 3.14. Main parameters of IR2153 are:

the maximum voltage at the VB terminal relative to the common wire is 600 V;
supply voltage (V cc) - 15 V;
consumption current (I cc) - 5 mA;
maximum control current I o -+100 mA / -210 mA;
turn-on time (t op) - 80 ns;
turn-off time (t off) - 40 ns;
switching pause (delay) -1.2 µs.

Rice. 3.14. Structural diagram of IC IR2153

Schematic diagram of the electronic ballast, made on the basis of IR2153, is shown in fig. 3.15.

IR2153 is a high power insulated gate field effect transistor (MOSFET) driver with an internal oscillator. It is an exact copy of the generator used in the 555 series timer, the domestic analogue is KR1006VI1. Operates directly from the DC bus through the quenching resistor R1.

Internal voltage regulation prevents Vcc from exceeding 15.6V. Undervoltage blocking blocks both gate drive outputs VT1 and VT2 when Vcc is below 9V.

DA1 has two control outputs:

lower 5 to control VT2;
the upper 7 output for controlling VT1, "floating", since the pulse shaper for controlling the field-effect transistor VT1 is powered by a floating power source, which is formed by the elements VD2, C7).

Rice. 3.15. Schematic diagram of an electronic ballast based on IR2153

When managing power keys(VT1, VT2), the IR2151 chip provides a switching delay of 1.2 µs to prevent the situation when transistors VT1 and VT2 are simultaneously open and through current flows through them, which instantly disables both transistors.

This ballast is designed to power one or two lamps with a power of 40 (36) W (lamp current - 0.43 A) from an alternating current network of 220 V 50 Hz. When using two lamps of 40 W, it is necessary to add the elements marked with a dotted line (EL2, L3, C11, RK3). It should be noted that for stable operation, the values of the elements in parallel branches must be equal (L3, C11 \u003d L2, C10), and the length of the wires supplied to the lamps must be the same.

Advice. When using one driver for two lamps, it is preferable to use frequency heating of the electrodes (without posistors). This method will be discussed below (when describing the electronic ballast on the IR53HD420 chip).

When using lamps of a different power (18-30 W), the values \u200b\u200bof L2 \u003d 1.8-1.5 mH should be changed (respectively); when using lamps with a power of 60-80 W - L2 \u003d 1-0.85 mH, a R2 - from the condition of fulfilling F g ~ F b (the formulas for calculating these frequencies are given below).

Mains voltage 220 V is supplied to network filter(electromagnetic compatibility filter) formed by elements C1, L1, C2, C3. The need for its use is due to the fact that key converters are sources of electromagnetic radio frequency interference, which network wires radiate into the surrounding space like antennas.

The current Russian and foreign standards regulate the levels of radio interference generated by these devices. Good results are obtained by two-section LC filters and screening of the entire structure.

At the input of the mains filter, a traditional unit for protecting against mains overvoltages and impulse noise is included, including a varistor RU1 and a fuse FU1. Thermistor RK1 with a negative temperature coefficient (NTC) limits the input current surge caused by the charge of the capacitive filter C4 at the input of the inverter when the electronic ballast is connected to the network.

Further, the mains voltage is rectified by the diode bridge VD1 and smoothed out by capacitors C4. The R1C5 chain feeds the DAI - IR2153 chip. The frequency of the internal oscillator FT of the microcircuit is set by the elements R2 = 15 kOhm; C6 \u003d 1 nF in accordance with the formula

The resonant frequency of the ballast circuit F6 is set by the elements L2 = 1.24 mH; C10 = 10 nF according to formula

To ensure good resonance, the following condition is required: the frequency of the internal generator should be approximately equal to the resonant frequency of the ballast circuit, i.e. Fg ~ Fb.

Construction and details. The mains filter inductor L1 is wound on a K32x20x6 M2000NM ferrite ring with a two-wire network wire until the window is completely filled. It is possible to replace the choke from the PFP power supply of a TV, VCR, computer.

Good noise suppression results are provided by specialized EPCOS filters: B8414-D-B30; B8410-B-A14.

The inductor of the electronic ballast L2 is made on a W-shaped magnetic core made of M2000NM ferrite. Core size Ш5х5 with gap 8 = 0.4 mm. The size of the gap in our case is the thickness of the gasket between the contacting surfaces of the halves of the magnetic circuit. It is possible to replace the magnetic circuit with Sh6x6 with a gap δ = 0.5 mm; Ш7х7 with a gap

δ = 0.8 mm.

To make a gap it is necessary to lay gaskets of non-magnetic material (non-foil fiberglass or getinax) of appropriate thickness between the mating surfaces of the halves of the magnetic circuit and fasten with epoxy glue.

The value of the inductance of the inductor (with a constant number of turns) depends on the value of the non-magnetic gap. With a decrease in the gap, the inductance increases, with an increase, it decreases. Reducing the gap is not recommended, because this leads to saturation of the core.

When the core is saturated, its relative magnetic permeability decreases sharply, which entails a proportional decrease in inductance. The decrease in inductance causes an accelerated increase in current through the inductor and its heating. The current passing through the LL also increases, which negatively affects its service life. The rapidly increasing current through the inductor also causes shock current overloads of power switches VT1, VT2, increased ohmic losses in the switches, their overheating and premature failure.

Winding L2- 143 turns of PEV-2 wire with a diameter of 0.25 mm. Interlayer insulation - varnished cloth. Winding - turn to turn. The main dimensions of W-shaped core c (consist of two identical W-shaped cores) of soft magnetic ferrites (according to GOST 18614-79) are given in Table. 3.2.

Table 3.2. Main dimensions of W-shaped cores

Transistors VT1, VT2 - IRF720, high power insulated gate field effect transistors. MOSFET is a Metal Oxide Semiconductor Field Effect Transistor; in the domestic version, MOSFETs are field-effect transistors of the metal-oxide-semiconductor structure.

Consider their parameters:

drain (ID) - 3.3 A;
pulsed drain current (I DM) -13 A;
maximum drain-source voltage (V DS) - 400 V;
maximum power dissipation (P D) - 50 W;
operating temperature range (Tj) - from -55 to +150 °С;
open resistance -1.8 Ohm;
total gate charge (Q G) - 20 nC;
input capacitance (C ISS) - 410 pF.

When choosing and replacing transistors(comparison in table 3.3) for electronic ballasts should be remembered that today the number of firms producing field-effect transistors is quite large (IR, STMicro, Toshiba, Fairchild, Infineon, etc.). The range of transistors is constantly expanding, more advanced ones with improved characteristics appear. Parameters to pay special attention to:

direct current drain (ID);
maximum drain-source voltage (VDS);
open resistance, RDS(on);
total gate charge (QG);
CISS input capacitance.

Possible replacement transistors for electronic ballast: IRF730, IRF820, IRFBC30A (International Rectifier); STP4NC50, STP4NB50, STP6NC50, STP6NB50 (STMicroelectronics); field-effect transistors from Infineon (http://www.infineon.com) series LightMos, CoolMOS, SPD03N60C3, ILD03E60, STP03NK60Z; PHX3N50E from PHILIPS, etc.

The transistors are mounted on small plate heatsinks. The length of the conductors between the driver outputs 5, 7, the resistors in the gate circuits R3, R4 and the gates of the field-effect transistors must be minimal.

Table 3.3. Comparison table with the parameters of some transistors for electronic ballasts

Rice. 3.16. The main dimensions of the core (to table. 3.2)

Diode bridge VD1 - imported RS207; permissible forward current 2 A; reverse voltage 1000 V. Can be replaced by four diodes with the appropriate parameters.

Diode VD2 class ultra-fast (superfast) - reverse voltage of at least 400 V; permissible direct direct current - 1 A; reverse recovery time - 35 ns. Fits 11DF4, BYV26B/C/D, HER156, HER157, HER105-HER108, HER205-HER208, SF18, SF28, SF106-SF109, BYT1-600. This diode should be located as close to the chip as possible.

Chip DAI - IR2153, it is interchangeable with IR2152, IR2151, IR2153D, IR21531, IR2154, IR2155, L6569, MC2151, MPIC2151. When using the IR2153D, the VD2 diode is not required, since it is installed inside the microcircuit.

Resistors R1-R5 - OMLT or MLT.

Capacitors C1-SZ - K73-17 for 630 V; C4 - electrolytic (imported) for a rated voltage of at least 350 V; C5 - electrolytic for 25 V; C6 - ceramic for 50 V; C7 - ceramic or K73-17 for a voltage of at least 60 V; C8, C9 - K73-17 for 400 V; SU - polypropylene K78-2 for 1600 6.

Varistor RU1 from EPCOS - S14K275, S20K275, replace with TVR (FNR) 14431, TVR (FNR) 20431 or domestic CH2-1a-430 V.

Thermistor (thermistor) RK1 with a negative temperature coefficient (NTC - Negative Temperature Coefficient) - SCK 105 (10 Ohm, 5 A) or EPCOS - B57234-S10-M, B57364-S100-M.

The thermistor can be replaced with a 4.7 ohm wirewound resistor with a power of 3-5 watts.

The RK2 posistor is a PTC thermistor (Positive Temperature Coefficient) with a positive temperature coefficient. The developers of IR2153 recommend using a posistor from Vishay Cera-Mite - 307C1260. His Main settings:

nominal resistance at +25 °С - 850 Ohm;
instantaneous (maximum allowable) rms voltage applied to the posistor when the lamp is ignited - 520 V;
constant (maximum allowable) rms voltage applied to the posistor during normal lamp operation, -175 V;
maximum allowable switching current (translating the posistor into a high-resistance state) -190 mA;
the diameter of the posistor is 7 mm.

A possible replacement for the RK2 posistor is EPCOS pulsed posistors (the number of switching cycles is 50000-100000): B59339-A1801-P20, B59339-A1501-P20, B59320-J120-A20, B59339-A1321-P20.

Posistors with the necessary parameters in an amount sufficient for eight electronic ballasts can be made from the widely used ST15-2-220 thesistor from the demagnetization system of the ZUSCT TV. After disassembling the plastic case, two "tablets" are removed. With a diamond file, two notches are made crosswise on each, as shown in fig. 3.17, and break it into four pieces along the cuts.

Advice. It is very difficult to solder leads to the metallized surfaces of a posistor made in this way. Therefore, as shown in Fig. 3.18, make a rectangular hole in the printed circuit board (pos. 3) and clamp the "tablet" fragment (pos. 1) between the elastic contacts (pos. 2) soldered to the printed conductors. By selecting the size of the fragment, you can achieve the desired duration of the lamp warm-up.

Rice. 3.17. "Tablet" posistor with notches

Rice. 3.18. Mounting a homemade posistor on the board

Advice. If the fluorescent lamp is supposed to be used in the infrequent on-off mode, then the posistor can be excluded.

Setting. The spread of the parameters of the elements C6, L2, SU may require adjustment of the driver frequency. The equality of the frequency of the master oscillator of the IR2153 microcircuit to the resonant frequency of the L2C10 circuit is easiest to achieve by selecting the frequency-setting resistor R2. To do this, it is convenient to temporarily replace it with a pair of series-connected resistors: constant (10-12 kOhm) and trimmer (10-15 kOhm). The criterion for the correct setting is reliable start-up (ignition) and stable burning of the lamp.

The ballast is assembled on a printed circuit board made of foil fiberglass and placed in an aluminum shielding casing. The printed circuit board and arrangement of elements is shown in fig. 3.19.

Rice. 3.19. Printed circuit board and layout of elements

So the first power supply, let's call it "high-voltage":

The second power supply, we will conditionally call it a "self-powered UPS":

Power supply number three, let's call it "powerful on 460x transistors" or simply "powerful 460":

And the last impulse for today is a "switching power supply for light bulbs", we will conditionally call it that.

PULSE POWER SUPPLY WITH YOUR HANDS ON IR2153

Functionally, the IR2153 microcircuits differ only in the diode installed in the planar package.

Functional diagram of IR2153

Functional diagram of IR2153D

CHIP	Maximum driver voltage	Start supply voltage	Stop supply voltage	Maximum current for driving the gates of power transistors / rise time	Maximum current for discharging the gates of power transistors / fall time	Internal zener voltage
				100 mA / 80...120 nS	210 mA / 40...70 nS
				NOT SPECIFIED / 80...150 nS	NOT SPECIFIED / 45...100 nS
				210 mA / 80...120 nS	420 mA / 40...70 nS

POPULAR TRANSISTORS FOR SWITCHED POWER SUPPLY
NAME	VOLTAGE	POWER	CAPACITY SHUTTER	Qg (MANUFACTURER)
NETWORK (220 V)
				17...23nC ( ST)
				38...50nC ( ST)
				35...40nC ( ST)
				39...50nC ( ST)
				46nC ( ST)
				50...70nC ( ST)
				75nC( ST)
				84nC ( ST)


				65nC ( ST)



				46nC ( ST)
				50...70nC ( ST)
				75nC( ST)
				65nC ( ST)
STP20NM60FP				54nC ( ST)






				150nC (IR) 75nC( ST)



				150...200nC (IN)
				252...320nC (IN)
				87...117nC ( ST)

I g \u003d Q g / t on \u003d 63 x 10 -9 / 120 x 10 -9 \u003d 0.525 (A) (1)

With the amplitude of the control voltage pulses at the gate Ug = 15 V, the sum of the output resistance of the driver and the resistance of the limiting resistor should not exceed:

R max = U g / I g = 15 / 0.525 = 29 (ohm) (2)

We calculate the output output impedance of the driver stage for the IR2155 chip:

R on \u003d U cc / I max \u003d 15V / 210mA \u003d 71.43 ohms
R off \u003d U cc / I max \u003d 15V / 420mA \u003d 33.71 ohms

To dispel doubts, the microcircuit was turned on and a two-channel oscilloscope was connected to it:

With the theory sorted out a bit, you can proceed to practical schemes.
The simplest IR2153 switching power supply circuit is an electronic transformer with a minimum of functions:

There are no additional functions in the circuit, and the secondary bipolar power supply is formed by two rectifiers with a midpoint and a pair of dual Schottky diodes. The capacitance of the capacitor C3 is determined on the basis of 1 microfarad of capacitance per 1 W of load. Capacitors C7 and C8 are of equal capacity and are located in the range from 1 uF to 2.2 uF. The power depends on the core used and the maximum current of the power transistors and theoretically can reach 1500 watts. However, this is only THEORETICALLY , assuming 155 VAC is applied to the transformer and the maximum current of the STP10NK60Z reaches 10A. In practice, in all datasheets, a decrease in the maximum current is indicated depending on the temperature of the transistor crystal, and for the STP10NK60Z transistor, the maximum current is 10 A at a crystal temperature of 25 degrees Celsius. At a crystal temperature of 100 degrees Celsius, the maximum current is already 5.7 A, and we are talking about the temperature of the crystal, and not the heat sink flange, and even more so about the temperature of the radiator.
Therefore, the maximum power should be selected based on the maximum current of the transistor divided by 3 if this is a power supply for a power amplifier and divided by 4 if this is a power supply for a constant load, such as incandescent lamps.
Given the above, we get that for a power amplifier you can get a switching power supply with a power of 10 / 3 \u003d 3.3A, 3.3A x 155V \u003d 511W. For a constant load, we get a power supply 10 / 4 \u003d 2.5 A, 2.5 A x 155V \u003d 387W. In both cases, 100% efficiency is used, which does not happen in nature.. In addition, if we proceed from the fact that 1 μF of the primary power capacitance per 1 W of load power, then we need a capacitor or capacitors with a capacity of 1500 μF, and such a capacitance already needs to be charged through soft start systems.
A switching power supply with overload protection and soft start for secondary power is shown in the following diagram:

Of course, the secondary winding counts on the voltage that is needed. The power supply can be soldered on the same board without mounting elements that are not on the diagram.

Well, in the end, several circuits of switching power supplies found on the Internet.
Scheme No. 6 is taken from the SOLDERING IRON website:

VIDEO INSTRUCTIONS FOR SELF-ASSEMBLY
PULSE POWER SUPPLY BASED ON IR2153 OR IR2155

A few words about the manufacture of pulse transformers:

How to determine the number of turns without knowing the brand of ferrite:

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